Power amplifier circuitry and method

ABSTRACT

A method and apparatus is provided for use in power amplifiers for reducing the peak voltage that transistors are subjected to. A power amplifier is provided with first and second switching devices and an inductor connected between the switching devices. The switching devices are driven such that the switching devices are turned on and off during the same time intervals.

CROSS-REFERENCE TO RELATED APPLICATIONS

The following U. S. patent applications filed concurrently herewith:Ser. No. ______, entitled “RF Power Amplifier Circuitry and Method forAmplifying RF Signals” by Timothy J. Dupuis et al; Ser. No. ______,entitled “Method and Apparatus for Regulating a Voltage” by Timothy J.Dupuis et al; Ser. No. ______, entitled “Dual Oxide Gate Device andMethod for Providing the Same” by Timothy J. Dupuis et al; Ser. No.______, entitled “RF Power Amplifier Device and Method for Packaging theSame” by Timothy J. Dupuis et al; and Ser. No. ______, entitled“Apparatus and Method for Providing Differential-to-Single Ended OutputConversion and Impedance Transformation” by Susanne A. Paul et al, areexpressly incorporated herein by reference.

FIELD OF THE INVENTION

This invention relates to the field of power amplifiers. Moreparticularly, this invention relates to improved power amplifiercircuitry which reduces the peak voltages to which switching devices ofthe amplifier are subjected.

BACKGROUND OF THE INVENTION

In some applications utilizing a power amplifier, it is desirable tolimit the peak voltage that the switching devices of the power amplifierare subjected to. For example, in CMOS devices, the transistor breakdownvoltage may be only slightly greater than the supply voltage. Therefore,CMOS devices are not well suited to traditional power amplifier designs,where switching devices are subjected to voltages at least twice thesupply voltage.

FIG. 1 is a schematic diagram of a conventional Class E amplifier. Asshown, a transistor M1 is connected between ground and an inductor L1which is connected to a voltage source V_(dd). The gate of thetransistor M1 is connected to an input signal Vi. The connection of thetransistor M1 and the inductor L1 forms a node labeled Vd. The switchingdevice M1, as well as other switching devices described may be comprisedof any suitable switching devices, for example, MOSFETs or othertransistor types. A capacitor C1 is connected between Vd and ground. Theamplifier includes a transformation network consisting of inductor L2and capacitor C2. The capacitor C2 is connected to a load R_(L) atoutput node V_(o).

FIG. 2 is a timing diagram illustrating the input signal Vi and theresulting voltage at Vd. As shown, the input signal Vi is a square wavesignal switching between ground and V_(dd). When the input signal Vi ishigh (V_(dd)), the transistor M1 is turned on, holding Vd to ground.When the input signal Vi transitions to low, transistor M1 turns off andthe voltage at Vd rises above V_(dd). During this time, the transistorM1 must sustain this high drain-to-source voltage. After peaking, thevoltage at Vd decreases until it reaches ground. In a typical prior artClass E design, this peak voltage is approximately 3.6 V_(dd). Althoughthe peak voltage can be reduced slightly, it can not be decreased belowabout 2.5 V_(dd) since the average voltage at Vd must equal V_(dd).Designs such as that shown in FIG. 1 are not well suited to certaindevice technologies, such as CMOS, where transistor breakdown voltagesare only slightly higher than the supply voltage.

It can therefore be seen that there is a need for amplifier designswhere the peak voltages applied to the transistors of the amplifier arereduced so that they are below the transistor breakdown voltages of thedevices being used to implement the design.

Another problem relating to amplifiers relates to the use ofdifferential circuits. It is difficult to performdifferential-to-single-ended conversion when a single ended load isrequired with high efficiency. Therefore, there is a need for improveddifferential-to-single-ended conversion designs.

SUMMARY OF THE INVENTION

A power amplifier of the invention includes a first switching deviceconnected between a first supply voltage and a first output node, asecond switching device connected between a second supply voltage and asecond output node, and an inductance coupled between the first andsecond output nodes.

Another embodiment of the invention provides a method of reducing thepeak output voltage in an amplifier including the steps of providing aninductor having first and second terminals, providing a first switchingdevice connected between the first terminal of the inductor and a firstsupply voltage, providing a second switching device connected betweenthe second terminal of the inductor and a second supply voltage,applying a voltage between the first and second terminals of theinductor during a first portion of a clock cycle by turning on the firstand second switching devices, and turning off the first and secondswitching devices during a second portion of the clock cycle.

Another embodiment of the invention provides a differential poweramplifier including a first amplifier having a first switching deviceconnected between a first supply voltage and a first output node, asecond switching device connected between a second supply voltage and asecond output node, and an inductance coupled between the first andsecond output nodes, a second amplifier having a third switching deviceconnected between a third supply voltage and a third output node, afourth switching device connected between a fourth supply voltage and afourth output node, and an inductance coupled between the third andfourth output nodes, and wherein the first and second amplifiers arecoupled together to drive a load.

Other objects, features, and advantages of the present invention will beapparent from the accompanying drawings and from the detaileddescription that follows below.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example and notlimitation in the figures of the accompanying drawings, in which likereferences indicate similar elements and in which:

FIG. 1 is a schematic diagram of a prior art Class E amplifier.

FIG. 2 is a timing diagram illustrating the voltage at V_(D) relative tothe input signal V₁ for the prior art Class E amplifier shown in FIG. 1.

FIG. 3 is a block diagram illustrating an example of an environment inwhich a power amplifier of the present invention may be used.

FIG. 4 is a schematic diagram of one embodiment of a power amplifier ofthe present invention.

FIG. 5 is a timing diagram illustrating the voltages present in theamplifier shown in FIG. 4, relative to the input signals.

FIG. 6 is a schematic diagram of an embodiment of a power amplifier ofthe present invention with a load connected differentially.

FIG. 7 is a schematic diagram of an embodiment of a power amplifier ofthe present invention connected to a single-ended load.

FIG. 8 is a schematic diagram of an embodiment of a power amplifier ofthe present invention connected differentially.

FIG. 9 is a timing diagram illustrating the voltages present in theamplifier shown in FIG. 8.

FIG. 10 is a schematic diagram of an embodiment of a power amplifier ofthe present invention.

FIG. 11 is a schematic diagram of another embodiment of a poweramplifier of the present invention.

FIG. 12 is a schematic diagram of an embodiment of a power amplifier ofthe present invention having a preamplifier circuit.

FIG. 13 is a timing diagram illustrating the voltages present in theamplifier shown in FIG. 12.

FIG. 14 is a schematic diagram of an embodiment of a two-stagedifferential power amplifier of the present invention.

FIG. 15 is a schematic diagram of a prior art circuit used forperforming differential-to-single-ended conversion.

FIG. 16 is a block diagram of a differential-to-single-ended conversionand impedance transformation circuit of the present invention.

FIGS. 17 is a schematic diagram of a differential-to-single-endedconversion and impedance transformation circuit of the presentinvention.

FIGS. 18 and 19 are schematic diagrams illustrating differential inputsAC-coupled from a load.

FIG. 20 is a block diagram of a differential-to-single-ended conversionand impedance transformation circuit having multiple differentialinputs.

FIG. 21 is a block diagram of a voltage regulator of the presentinvention.

FIG. 22 is a schematic diagram of an embodiment of a voltage regulatorof the present invention.

FIG. 23 is a schematic diagram of an embodiment of a voltage regulatorof the present invention.

FIG. 24 is a schematic diagram of an embodiment of a voltage regulatorof the present invention.

FIG. 25 is an isometric view illustrating how a device of the presentinvention is packaged.

FIG. 26 is a side view of the device shown in FIG. 25.

FIG. 27 is a diagram illustrating a ceramic chip carrier with aninductor formed in the carrier.

FIG. 28 is a diagram illustrating a ceramic chip carrier with avertically-formed inductor formed in the carrier.

FIG. 29 is an electrical schematic diagram of inductors connectedbetween four connection points.

FIG. 30 is a diagram illustrating an example of how the inductors shownin FIG. 29 could be formed in a ceramic chip carrier.

DETAILED DESCRIPTION

In order to provide a context for understanding this description, thefollowing illustrates a typical application of the present invention. Apower amplifier of the present invention may be used as an amplifier foruse with a wireless transmission system such as a wireless telephone orother device. The invention may also be applied to other applications,including, but not limited to, RF power amplifiers. In a wireless devicesuch as a cellular telephone, the device may include a transceiver, anantenna duplexer, and an antenna. Connected between the transceiver andthe antenna duplexer is an RF power amplifier for amplifying signals fortransmission via the antenna. This is one example of an application of apower amplifier of the present invention. Of course the invention may beused in any other application requiring a power amplifier. In the caseof a wireless telephone application, the invention may be applied to GSMor other constant envelope modulation systems.

FIG. 3 is a block diagram illustrating an example of an environment inwhich a power amplifier of the present invention may be used. FIG. 3shows a power amplifier 310 connected to a pair of input signals V_(in)and V_(ip). The input signals come from an input 312 from an inputnetwork such as the transceiver chip mentioned above. An input buffer isformed by a plurality of inverters X1 and X2 which are connected to theinput 312 as shown. The input buffer circuit could also be comprised ofmore or less inverters, or any other suitable circuitry. The poweramplifier 310 is also connected to a voltage regulator 314 whichprovides a regulated voltage source V_(dd) from a voltage source, suchas battery voltage VB. The power amplifier 310 is also connected to atransformation network 316 which is connected to a load 318. Note thatthe connection between power amplifier 310 and the transformationnetwork 316 may be comprised of a single or multiple connections. FIG. 3is shown with n connections. In the example of a wireless transmissionsystem, the load 318 may be comprised of an antenna. Note that thecomponents shown in FIG. 3 are optional and are not essential to thepower amplifier 310.

FIG. 4 is a schematic diagram of one embodiment of a power amplifier ofthe present invention. The power amplifier includes a switching deviceM1 connected between ground and the node labeled V_(dn). The gate of theswitching device M1 is connected to the input signal V_(in). Anotherswitching device M2 is connected between the voltage source V_(dd) and anode labeled V_(dp). The gate of the switching device M2 is connected tothe input signal V_(ip). Connected between the switching devices M2 andM1 is an inductor L1. FIG. 4 also shows a capacitor C1 connected betweenV_(dn) and ground. A capacitor C3 is connected between V_(dp) and Vdd.The capacitors C1 and C3 may be comprised of a combination of separatecapacitors and parasitic capacitances of the switching devices M1 andM2. The power amplifier shown in FIG. 4 also includes a reactive networkconnected between V_(dn) and the amplifier output V_(o). The reactivenetwork is formed by inductor L2 and capacitor C2 and can be used forfiltering or impedance transformation. A load R_(L) is connected to theamplifier output V_(o).

The power amplifier shown in FIG. 4 resembles a push-pull amplifiertopologically, but is fundamentally different, in that the input signalsV_(in) and V_(ip) are inverses of one another. Since switching device M1is an n-channel device and switching device M2 is a p-channel device,the switching devices M1 and M2 are both turned on and turned off duringthe same time intervals. FIG. 5 is a timing diagram illustrating thevoltages present in the amplifier shown in FIG. 4, relative to the inputsignals. FIG. 5 shows the input signals V_(in) and V_(ip) which are 180°out of phase with each other. In other words, when one of the inputsignals is high, the other is low. During phase 1 (V_(in) high andV_(ip) low), the switching devices M1 and M2 are both turned on so thatV_(dp) and V_(dn) are clamped to V_(dd) and ground respectively. Duringphase 2 (V_(in) low and V_(ip) high), the switching devices M1 and M2are both turned off. The voltage at V_(dn) rises and begins to ring at afrequency determined by the values of the components L1, C1, C3, L2, andC2. For the best efficiency, these components are chosen so that V_(dn)rises and then returns to ground immediately before the end of phase 2.The voltage at V_(dp) falls and rings in a similar way. The voltage atnode V_(dp) rises back to V_(dd) immediately before the end of phase 2,when switching devices M1 and M2 are turned on.

The peak voltages present across the switching devices M1 and M2 can beadjusted as desired by changing the passive component values in thecircuit under the constraint that the average voltage of V_(dn) mustequal that of V_(dp). If this average voltage lies at V_(dd)/2 then thepeak value of V_(dn) will be only slightly higher than V_(dd) and thatof V_(dp) will be only slightly lower than ground. The duty cycle of theinput signals V_(in) and V_(ip) waveforms can be adjusted to reduce thepeak voltages even further. As a result, this configuration eliminatesthe large signal swings that transistors are subjected to in the priorart.

The power amplifier shown in FIG. 4 does not take full advantage of thesignal swing that occurs on node V_(dp). An increase in efficiency canbe achieved by making use of the signal swing on both V_(dp) and V_(dn).This can be accomplished by connecting the load differentially acrossnodes V_(dp) and V_(dn) as shown in FIG. 6. FIG. 6 shows a poweramplifier similar to that shown in FIG. 4. The power amplifier includesswitching devices M1 and M2, inductor L1, and capacitors C1 and C3. Atransformation network 616 is connected to both nodes V_(dp) and V_(dn).A load R_(L) is connected to the transformation network 616. Thewaveforms for the power amplifier shown in FIG. 6 are similar to thosefor the power amplifier shown in FIG. 4. In this embodiment, the currentflowing through the load R_(L) is determined by the difference betweenthe voltages on V_(dp) and V_(dn).

When a single-ended load is required, the transformation network can bemade to facilitate a single-ended load. FIG. 7 shows a power amplifierwith two capacitors C2 and C4 and an inductor L3 connected as shownbetween V_(dn) and V_(o). An inductor L2 is connected between V_(dp) andthe connection point of the capacitors C2 and C4. A single-ended loadR_(L) is connected between V_(o) and ground. The waveforms for the poweramplifier shown in FIG. 7 are similar to those for the power amplifiershown in FIG. 4. In this embodiment, the current flowing to the outputfrom V_(dp) and current flowing to the output from V_(dn) add when theyare summed in phase at the load. The load is AC coupled from eitherV_(dp) or V_(dn) by capacitor C4. The inductor L2 and capacitor C2 canalso be chosen to transform the load impedance R_(L) into a desiredimpedance so that power delivered to the load can be adjustedindependently from the voltage swing on V_(dp) and V_(dn). In this case,the voltage swing on V_(o) will vary from that on V_(dp) and V_(dn) asdetermined by the selection of C2 and L2. Since the combination of L2and C2 is a tuned circuit, it provides some bandpass filtering. Ifadditional filtering is desired, capacitor C4 and inductor L3 can alsobe used as an additional bandpass filter. In summary, L2 and C2 in theconfiguration of FIG. 7 simultaneously perform the functions ofimpedance transformation, filtering, and differential-to-single-endedconversion.

The amplifier of the present invention may also be implementeddifferentially using two amplifiers (such as the amplifier shown in FIG.7) connected together as shown in FIG. 8. FIG. 8 shows a first amplifier(the positive side) comprised of switching devices M1+ and M2+, inductorL1+, capacitors C1+ and C3+, and a transformation network comprised ofcapacitors C2+ and C4+ and inductors L2+ and L3. A second amplifier (thenegative side) is comprised of switching devices M1− and M2−, inductorL1−, capacitors C1− and C3−, and a transformation network comprised ofcapacitors C2− and C4− and inductors L2− and L3. The two amplifiers aresimilar to each other with the inductors L2 and capacitors C2interchanged as shown. The input signals V_(in−) and V_(ip−) on thenegative side are shifted by 180 degrees from the input signals V_(in+)and V_(ip+) on the positive side. FIG. 9 is a timing diagramillustrating the voltages present at the nodes V_(dn+), V_(dp+),V_(dn−), and V_(dp−).

The values of the passive components in the amplifier shown in FIG. 8may be chosen so that the resulting currents from both amplifiers sum inphase at the load R_(L). The advantages of the power amplifier shown inFIG. 8 are similar to the advantages common to differential circuits ingeneral. For example, undesired interference from supply or substratenoise is common-mode. Another advantage is that the impact of supplyresistance is reduced because the supply current flows during both clockphases.

Note that the load R_(L) shown in FIG. 8 could be connected to only twoof the four output nodes of the power amplifier. For example, aconfiguration similar to that shown in FIG. 4 could be connecteddifferentially to the load R_(L), where the nodes V_(dp+) and V_(dp−)are not connected to V_(o).

FIG. 8 also shows an alternate embodiment where an optional inductor L4is connected (shown in dashed lines) between nodes V_(dp+) and V_(dp−).Without the optional inductor L4, the voltage swings on nodes V_(dp+),V_(dp−), V_(dn+) and V_(dn−) and the values of capacitors C1+, C1−, C3+and C3− can not be independently adjusted. The optional inductor L4 hasthe advantage that these voltage swings can be adjusted independently ofthe capacitance values mentioned above.

The capacitors C1 and C3 described above are used to shape the waveformsof the voltages on V_(dp) and V_(dn). As mentioned above, thesecapacitances may be provided by separate capacitors or by the parasiticcapacitances of switching devices M1 and M2. In another embodiment,these capacitances are formed by switching devices in a way thatimproves the efficiency of the amplifier.

FIG. 10 is a schematic diagram of a power amplifier similar to theamplifier shown in FIG. 8. In the amplifier shown in FIG. 10, thecapacitors C1+ and C3+ are replaced by switching devices M3− and M4−,respectively. Similarly, the capacitors C1− and C3− are replaced byswitching devices M3+ and M4+, respectively. Each of the switchingdevices M3 and M4 are driven as shown by a voltage from the oppositeamplifier. For example, the switching device M4+ is driven by thevoltage at V_(dp−) on the negative side. The switching device M4− isdriven by the voltage at V_(dp+) on the positive side. Similarly, theswitching device M3+ is driven by the voltage at V_(dn) while theswitching device M3− is driven by the voltage at V_(dn+). The waveformsfor the amplifier shown in FIG. 10 are similar to those described above.

The amplifier shown in FIG. 10 allows the switching devices M1+ and M1−to be made smaller by an amount equal to the size of switching devicesM3+ and M3−. Similarly, the switching devices M2+ and M2− can be madesmaller by an amount equal to the size of switching devices M4+ and M4−.However, switching devices M1 and M2 should remain sufficiently large toassure stability of the circuit. A decrease in the size of the switchingdevices M1 and M2 improves the efficiency since the input capacitancesthat must be driven are smaller. Another advantage to the amplifiershown in FIG. 10 is that cross-coupling helps to assure that thewaveforms present at V_(dp−) and V_(dn−) have the correct phaserelationship to the waveforms present at V_(dp+) and V_(dn+), despitepossible timing variations on the positive inputs (V_(ip+), V_(in+)) andon the negative inputs (V_(ip−), V_(in−)).

FIG. 10 also shows an alternate embodiment where an optional inductor L4is connected (shown in dashed lines) between nodes V_(dp+) and V_(dp−),similar to the inductor L4 shown in FIG. 8. If the optional inductor L4is connected, the voltage swings of nodes Vdp+, Vdp−, Vdn+, and Vdn− canbe chosen independently from the input capacitances of M4−, M4+, M3−,M3+.

FIG. 11 is a schematic diagram of a power amplifier similar to theamplifier shown in FIG. 10, but with the inductors L1+ and L1− replaced.Inductor L1+ is replaced with a pair of inductors L1A+ and L1B+.Inductor L1− is replaced with a pair of inductors L1A− and L1B−. Thenode formed by the connection of inductors L1A+ and L1B+ is connected tothe node formed by the connection of inductors L1A− and L1B−. Theembodiment shown in FIG. 11 has similar advantages to the embodiment inFIG. 10 with the optional inductor L4 in that it allows the voltageswings of nodes Vdp+, Vdp−, Vdn+, and Vdn− to be chosen independentlyfrom the input capacitances of M4−, M4+, M3−, M3+.

As described above with respect to FIG. 3, input buffer circuitry may beused to drive the gates of the switching devices M1 and M2 of theamplifiers described above. However, the efficiency may be improved if asimilar amplifier circuit is used as a preamplifier circuit. FIG. 12 isan example of an amplifier having a preamplifier circuit.

FIG. 12 shows an amplifier similar to the amplifier shown in FIG. 7. Atthe input of the amplifier, a preamplifier is shown. The preamplifier iscomprised of switching devices M5 and M6 connected between ground andV_(dd). An inductor L3 is connected between the switching devices M5 andM6. The preamplifier includes inputs V_(ip2) and V_(in2). Thepreamplifier circuit receives input signals V_(ip2) and V_(in2) andgenerates signals V_(ip) and V_(in) for use by the amplifier. Thepreamplifier circuit is similar to the amplifiers described above,except that all of the passive elements except inductor L3 areeliminated. The capacitances required by the preamplifier circuitry areformed from the input capacitances of the gates of switching devices M1and M2. Of course, other passive elements could be used with thepreamplifier circuit.

FIG. 13 is a timing diagram illustrating the waveforms at V_(in),V_(ip), V_(dn), and V_(dp) of FIG. 12. The preamplifier output waveformsV_(ip) and V_(in) have a shape that makes them well suited for drivingthe input gates of switching devices M1 and M2 in the final stage.

Note that in an alternate configuration the capacitor C4 could beconnected between inductor L2 and V_(o) with capacitor C2 connectedbetween V_(dn) and V_(o). This alternate configuration functionssimilarly to the configuration shown in FIG. 12.

FIG. 14 is a schematic diagram of an amplifier using a two-stagedifferential configuration which provides an increased efficiency overthe circuit shown in FIG. 12. The amplifier shown in FIG. 14 is similarto the differential amplifier shown in FIG. 10, with the addition ofpreamplifier circuitry. The inputs V_(ip+) and V_(in+) of the amplifierare connected to preamplifier circuitry comprised of switching devicesM5+ and M6+. The switching devices M5+ and M6+ are connected betweenground and V_(dd), with an inductor L3+ connected between them.Capacitances are provided to nodes V_(dp2+) and V_(dn+) by switchingdevices M8+ and M7+, respectively. The negative side of the amplifier isconfigured in the same manner. The positive and negative sides of thepreamplifier circuitry are cross-coupled in the same way as theamplifier circuitry shown in FIG. 10 (described above). In thisconfiguration, the input capacitances of the NMOS and PMOS switchingdevices M1 and M2 of the power amplifier, the input capacitances of thepreamplifier switching devices M7 and M8, and the value of inductor L5can be adjusted so that the signals at V_(dp2) and V_(dn2) have thedesired peak amplitudes.

Another aspect of the present invention relates to a circuit and methodof providing differential-to-single ended output conversion andimpedance transformation from differential signals. Differentialcircuits have a number of advantages that are well known. For example,the impact from noise sources is reduced since these signals arecommon-mode (i.e., the positive and negative sides are effectedidentically). In addition, even-order harmonics are reduced because ofcircuit symmetry. Because of these and other advantages, a differentialconfiguration may be desirable even when the load is single-ended. If asingle-ended load is needed, circuitry for differential-to-single-endedconversion is needed.

One prior art method for performing differential-to-single-endedconversion at high frequency involves use of a transformer or balun.FIG. 15 shows a prior art circuit used for performingdifferential-to-single-ended conversion using a transformer T1. Theprimary side of the transformer T1 is connected to a first differentialinput V₊ and a second differential input V₃₁. The secondary side of thetransformer T1 is connected to ground and an output node V_(O). A loadZ_(L) is connected between ground and the output node V_(O). If thetransformer has a 1-to-1 turns ratio, then the differential signals V₊and V⁻ are translated into a signal having an amplitude of (V₊-V⁻)across the load Z_(L).

In some applications, impedance matching or impedance transformation isneeded to transform a given load impedance into a different impedanceseen by the driver. Impedance transformation can be accomplished, aspart of the differential-to-single ended conversion, using thetransformer circuit shown in FIG. 15 by adjusting the winding ratio ofthe transformer T1. However, the use of transformers fordifferential-to-single-ended conversion and impedance transformation hasdisadvantages. First, high quality transformers are larger and morecostly than other passive elements and are not easily integrated withother semiconductor circuits. Second, practical transformers haveimperfect magnetic coupling which causes a loss of power from input tooutput.

The present invention provides a technique that performsdifferential-to-single ended conversion as well as impedancetransformation and avoids the disadvantages of a transformer solution.FIG. 16 shows a block diagram of a differential-to-single-endedconversion and impedance transformation circuit. The circuit has a firstimpedance X₁ coupled between the second differential input signal V⁻ andan output node V_(O). A second impedance X₂ is coupled between the firstdifferential input signal V₊ and the output node V_(O). A load Z_(L) isconnected between the output node V_(O) and ground. In the circuit shownin FIG. 16, current flowing to the output node V_(O) from differentialinput V₊ is shifted in phase from the voltage on V₊. Similarly, currentflowing to the output node V_(O) from differential input V⁻ is shiftedin phase from the voltage on V⁻. The impedances X1 and X2 are chosen sothat these two currents add together when they are summed at the loadZ_(L). For example, if X1 shifts the output current by +90 degrees andX2 shifts the output current by −90 degrees then the resultant currentswill sum in phase at the load. FIG. 17 illustrates one example of animplementation of the circuit shown in FIG. 16. FIG. 17 shows an L-Cdifferential-to-single-ended conversion and impedance transformationcircuit. The impedance X1 is comprised of a capacitor C5 which iscoupled between the second differential input signal V⁻ and the outputnode V_(O). The impedance X2 is comprised of an inductor L6 which iscoupled between the first differential input signal V₊ and the outputnode V_(O).

Referring back to FIG. 16, since the inputs V₊ and V⁻ are differential,the inputs have opposite signs. However, the differential inputs V₊ andV⁻ are not necessarily equal in amplitude. The output voltage V_(O) ofthe differential-to-single-ended conversion and impedance transformationcircuit is given by the following equation: $\begin{matrix}{V_{0} = {\frac{( {{V_{+}X_{1}} + {V_{-}X_{2}}} )( {{{- {jX}_{2}}X_{1}} + {( {X_{1} + X_{2}} )Z_{L}}} )}{( {( {X_{1}X_{2}} )^{2} + {( {X_{1} + X_{2}} )^{2}Z_{L}^{2}}} )}{Z_{L}.}}} & (1)\end{matrix}$The power P_(L) delivered to the load Z_(L) is given by the followingequation: $\begin{matrix}{P_{L} = {\frac{( {{V_{+}X_{1}} + {V_{\_}X_{2}}} )^{2}}{( {( {X_{1}X_{2}} )^{2} + {( {X_{1} + X_{2}} )^{2}Z_{L}^{2}}} )}{Z_{L}.}}} & (2)\end{matrix}$Differential-to-single-ended conversion is achieved if the impedances X₁and X₂ have opposite signs. Impedances X₁ and X₂ may be comprised of anycombination of reactive elements (e.g., capacitor C5 and inductor L6shown in FIG. 17) whose combination meets this requirement. For example,if differential inputs V₊ and V⁻ have equal amplitudes A, and impedancesX₁ and X₂ have equal amplitudes X, then the output voltage V_(O) can bedetermined by substituting these values into equation (1) above. Theresulting output voltage V_(O) is given by the following equation:$\begin{matrix}{V_{0} = {{- {j2A}}{\frac{Z_{L}}{X}.}}} & (3)\end{matrix}$

It can be seen from equation (3) that the ratio R/X can be chosen sothat the amplitude of the output V_(O) is either larger or smaller thanthe amplitude A of the differential input. The voltage of the outputV_(O) increases as the value of X decreases. Similarly, the voltage ofthe output V_(O) decreases as the value of X increases.

In certain applications, the load Z_(L) must be AC-coupled from one ofthe differential inputs V⁻ or V+. FIGS. 18 and 19 show examples of a howthe differential inputs may be AC-coupled from the load Z_(L) in theexample shown in FIG. 17. In the circuit shown in FIG. 18, an additionalcapacitor C6 is inserted between the output node V_(O) and both thecapacitor C5 and the inductor L6. The capacitor C6 AC-couples the outputnode V_(O) from the first and second differential inputs V₊ and V⁻. Inthe circuit shown in FIG. 19, an additional capacitor C6 is insertedbetween the output node V_(O) and the inductor L6. The capacitor C6AC-couples the output node V_(O) from the first differential input V₊.Note that the capacitor C1 provides AC-coupling between the output nodeV_(O) from the second differential input V⁻.

The techniques for providing differential-to-single-ended conversion andimpedance transformation described above can be applied to circuitshaving multiple differential inputs. FIG. 20 shows adifferential-to-single-ended conversion and impedance transformationcircuit having multiple differential inputs. FIG. 20 shows differentialinputs V₁ through V_(N), where N is the total number of differentialinputs. A first impedance X₁ is coupled between the differential inputV₁ and the output node V_(O). A second impedance X₂ is coupled betweenthe differential input V₁ and the output node V_(O). Similarly, an Nthimpedance X_(N) is coupled between the differential input V_(N) and theoutput node V_(O). Each of the currents from each differential input issummed in phase at the output node V_(O). In this embodiment, theimpedance X_(j) between the jth differential input V_(j) and the outputnode V_(O) will depend on its phase with respect to that of otherdifferential inputs. Optimal power transfer to the load Z₁ occurs whenthe impedances X_(j) are purely reactive. However, this technique maystill be applied when impedance X_(j) is not purely reactive. Forexample, this might occur when actual inductors and capacitors have aseries resistance.

As mentioned above, the RF power amplifier shown in FIG. 3 includes avoltage regulator 314 connected between the power amplifier 310 and asource of battery voltage VB to provide a voltage source VDD. In oneembodiment of the present invention, the voltage regulator 314 resideson the same integrated circuit as the power amplifier circuit. Thefunction of the voltage regulator is to provide a source of voltage tothe power amplifier and to help control the output power level. Forexample, in a cellular phone environment, a base station may dictate thepower level at which each cell phone should transmit (based on factorssuch as the physical distance from the base station, for example).Varying the voltage level (VDD) can control the output power of thepower amplifier. As the voltage of the voltage source VDD increases, theoutput power increases. Therefore, by controlling the operation of thevoltage regulator, and therefore controlling the voltage of voltagesource VDD, the output power of the amplifier can be controlled. Whilethe power amplifier 310 will function with any suitable voltageregulator or voltage source, described below is a detailed descriptionof a suitable voltage regulator.

FIG. 21 is a block diagram of a voltage regulator 544 used to provide aregulated voltage VDD from a voltage source VB, for example, from abattery. As shown, the regulated voltage VDD is provided to a device530. The device 530 may be any type of device requiring a voltage sourceincluding, but not limited to power amplifiers. The voltage regulator544 includes an input 546 that is connected to a control signal VSET tocontrol the voltage level VDD provided to the device 530. Following is adetailed description of the voltage regulator of the present inventionin the context of its use in an RF power amplifier (such as that shownin FIG. 3). However, it is understood that the voltage regulator may beused with any type of amplifier as well as any other type of devicerequiring a voltage source.

FIG. 22 is a schematic diagram of a first embodiment of a voltageregulator 644 connected to a battery voltage VB. The voltage regulator644 is comprised of a device M9 and an op amp X4. The op amp X4 includesa first input 646 for connection to a voltage control signal VSET. In apreferred embodiment, the control signal VSET is an analog voltagesignal that is proportional to the desired voltage level. The otherinput to the op amp X4 is connected to the regulated voltage VDD. Theoutput of the op amp X4 is connected to the input of the device M9.

FIG. 23 is a schematic diagram of another embodiment of a voltageregulator 744 connected to a battery voltage VB. The voltage regulator744 is similar to the voltage regulator 644 shown in FIG. 22 with theaddition of a second regulator circuit comprised of op amp X5, switchingdevice M10, and an external resistor R1. FIG. 23 also shows anintegrated circuit 770 (dashed lines) to illustrate that the poweramplifier is formed on the integrated circuit 770 while the resistor R1is not. The integrated circuit 770 may also be the same integratedcircuit on which the device to be powered resides.

The first regulator circuit is connected in the same manner as theregulator circuit shown in FIG. 22. The op amp X5 of the secondregulator circuit includes an input VSET2 for connection to a voltagecontrol signal. The other input to the op amp X5 is connected to theregulated voltage VDD. The output of the op amp X5 is connected to thegate of the device M10. The external resistor R1 is connected betweenthe battery voltage VB and the device M10. FIG. 23 also shows voltagecontrol circuitry 776 which has an input 746 connected to the controlsignal VSET. The voltage control circuitry 776 uses the signal VSET tocreate voltage control signals VSET1 and VSET2 for use by the first andsecond regulator circuits. By controlling both regulators, the voltagelevel VDD can be controlled. In addition, by selectively activating thesecond regulator, power can be dissipated off the integrated circuit 770(via resistor R1). This results in a reduction of heat generated in theintegrated circuit 770.

The voltage regulator 744 operates as follows. Since it is desired tominimize the amount of power dissipated on the integrated circuit 770,one goal is to maximize the use of the second regulator circuit (X5,M10) in order to maximize power dissipation through the externalresistor R1. Therefore, voltage control circuitry 776 will enable thesecond regulator circuit to provide as much power as it can beforeenabling the first regulator circuit (X4, M9). In other words, when morepower is required than the second regulator circuit can provide, thefirst regulator circuit is enabled to provide additional power. In thisway, the maximum amount of power will be dissipated through externalresistor R1.

FIG. 24 is a schematic diagram of another embodiment of voltageregulator 844 having multiple regulators and multiple externalresistors. The voltage regulator 844 is similar to the regulator 744shown in FIG. 23, with the addition of a third regulator circuitcomprised of device M11, op amp X6, and external resistor R2. The thirdregulator circuit is connected in the same ways as the second regulatorcircuit, and operates in a similar manner. The op amp X6 of the thirdregulator circuit includes an input VSET3 for connection to a voltagecontrol signal. The other input to the op amp X5 is connected to theregulated voltage VDD. The output of the op amp X6 is connected to thegate of device M11. The external resistor R2 is connected between thebatter voltage VB and device M11. FIG. 24 also shows voltage controlcircuitry 876 which has an input 846 connected to the control signalVSET. The voltage control circuitry 876 uses the signal VSET to createvoltage control signals VSET1, VSET2, and VSET3 for use by the regulatorcircuits. By activating the second or third regulator, power can bedissipated off the integrated circuit 870 (via resistor R1 and/or R2).This results in a reduction of heat generated in the integrated circuit870.

The voltage regulator 844 operates as follows. Since it is desired tominimize the amount of power dissipated on the integrated circuit 870,one goal is to maximize the use of the second and third regulatorcircuits in order to maximize power dissipation through the externalresistors R1 and R2. Therefore, voltage control circuitry 876 willenable the second and third regulator circuits to provide as much poweras it can before enabling the first regulator circuit. In other words,when more power is required than the second and/or third regulatorcircuit can provide, the first regulator circuit is enabled to provideadditional power. In this way, the maximum amount of power will bedissipated through external resistors R1 and R2.

The values of the resistors R1 and R2 may be equal, or may be different,depending on the needs of a user. In addition, the invention is notlimited to the use of one or two external resistors. Additionalregulator circuits and external resistors could be added. In oneembodiment, the value of resistor R1 is 0.7 ohms and the value ofresistor R2 is 0.3 ohms.

Another benefit of the present invention involves the use of dual gateoxide devices. In CMOS digital systems, it is sometimes desired toprovide devices suitable for use with two voltage levels (e.g., 3.3volts and 5 volts). Therefore, processing technologies have beendeveloped to provide a single integrated circuit having both 0.5 μm and0.35 μm devices. As mentioned above, a thicker gate oxide results in adevice with a higher breakdown voltage. On the other hand, a thinnergate oxide results in a faster device, but with a lower breakdownvoltage.

The RF amplifier of the present invention takes advantage of theavailability of dual gate oxide devices by selectively choosing certaingate lengths for various components of the amplifier. For example, ithas been discovered that for preprocessing circuitry or pre-drivercircuitry, a high speed is desirable and breakdown voltage is not asimportant. Therefore these devices are designed using a thinner gateoxide. For output state devices, where a high breakdown voltage is moreimportant, the devices are designed using a thicker gate oxide.

In one embodiment, a dual gate oxide device is used to create an RFamplifier such as the RF amplifier shown in FIGS. 12, and 14. Onesuitable use of dual gate oxides in these amplifiers is to utilizedevices having channel lengths of both 0.5 μm and 0.35 μm. The 0.5 μmand 0.35 μm devices have gate oxide thicknesses of 140 Angstroms (Å) and70 Å, respectively. Referring to the example shown in FIG. 12, thepredriver devices M5 and M6 can be chosen with much smaller devicewidths than the output devices M1 and M2. In this case, the predriveroutput signals Vip and Vin are nearly sinusoidal, the voltage difference(Vip−Vin) varies between about +Vdd and −Vdd, and the input capacitancesof M1 and M2 can be chosen so that neither M5 nor M6 experiences avoltage drop that is larger than Vdd. As a result, a high breakdownvoltage is not critical for the predriver and devices M5 and M6 can beimplemented using 0.35 μm devices. When high efficiency is desired,switching devices M1 and M2 of the final amplifier stage are sized withlarge device widths so that nodes Vdn and Vdp are strongly clamped totheir respective supply voltages of ground and Vdd when these devicesare on. In this case, the voltage difference (Vdp−Vdn) varies over arange that is larger than that of the predriver and either M1, M2, orboth will experience a voltage drop that is larger than Vdd. Since ahigher breakdown voltage is desired from these devices, M1 and M2 caneach be implemented using 0.5 μm devices. Since PMOS transistors aretypically slower than NMOS transistors and thicker gate oxide devicesare slower than thinner gate oxide devices, it is preferable to use athicker gate oxide for NMOS devices than for PMOS devices. An example ofthe use of dual gate oxide thicknesses for the RF amplifier of FIG. 14includes only NMOS devices with a thick gate oxide. Predrivertransistors M5+, M5−, M6+, M6−, M7+, M7−, M8+, and M8− are implementedusing using 0.35 μm devices because, as described above, they are notsubjected to voltage drops greater than Vdd and breakdown is not acritical concern. As described above, the final amplifier stageexperiences larger voltage swings. However these larger swings can bedistributed across across its NMOS and PMOS devices in such a way thatonly NMOS devices see a voltage swing larger than Vdd. This isaccomplished by adjusting the values of inductors L1+, L1−, and L4 andthe input capacitances of devices M3+, M3−, M4+, and M4−. In thisapproach, PMOS devices M2+, M2−, M4+, and M4− in the final amplifierstage are thinner gate oxide devices, whereas NMOS devices M1+, M1−,M3+, M3− are thicker gate oxide devices.

Of course, the present invention is not limited to the values describedabove. For example, as thinner gate oxides become more common, one orboth thicknesses may become lower. In addition, note that the terms“thicker” or “thinner” in this description are intended to only refer tointentional or significant differences in gate oxide thicknesses. Forexample, the 0.35 μm devices may vary from one another by some smallamount depending on manufacturing tolerances. A 0.5 μm device isconsidered to be “thicker” than a 0.35 μm device. Also note that thisinvention applies to various CMOS devices and that the RF Amplifierdescribed above is only used as one example of the application of dualgate oxide devices of the present invention.

Another benefit of the present invention relates to how an RF poweramplifier of the present invention is packaged. The design of an RFamplifier requires a low inductance and low resistance to thetransistors or switching devices. In addition, RF power amplifierdesigns typically require a number of passive components such asinductors and capacitors. It is advantageous to integrate thesecomponents in the power amplifier package. The packaging technique ofthe present invention addresses these concerns by using “flip chip”technology and multi-layer ceramic chip carrier technology.

FIGS. 25 and 26 are isometric and side views, respectively, illustratinga packaging technique of the present invention. FIGS. 25 and 26 show a“flip chip” integrated circuit 970 mounted to a multi-layer ceramic chipcarrier 972. The integrated circuit 970 includes a plurality ofconnection points, or “bumps” 974 on the underside of the integratedcircuit 970. Similarly, the ceramic chip carrier 972 includes aplurality of connection points or bumps 976. The bumps 974 of theintegrated circuit 970 are formed by solder and can be mounted tocorresponding conductive material formed on the upper surface of theceramic chip carrier 972. Similarly, the bumps 976 of the ceramic chipcarrier 972 are also formed by solder and are used to mount the chipcarrier 972 to a printed circuit board (not shown). A typical flip chipallows 250 μm spaced bumps. A typical chip carrier also allows 250 μmspaced vias for connection to the flip chip bumps 974. In one example,6×6 mm ceramic chip carrier includes 36 bumps 976 for connection to aPCB. Flip chip and ceramic chip carrier technologies are consideredconventional and will not be described in detail.

Various benefits can be realized by selectively placing certaincomponents of the RF power amplifier of the present invention onintegrated circuit 970 and ceramic chip carrier 972. The invention willbe described with respect to the RF power amplifier shown in FIG. 14,although the invention is not limited to power amplifiers. In oneembodiment of the invention, all of the switching devices are formed onthe integrated circuit 970. In addition, the power transistors (such asswitching devices M1+, M1−, M2+, M2−) formed on the integrated circuit970 are preferably placed directly below the bumps 974 of the integratedcircuit 970 resulting in low resistance and inductance (as compared towire bond integrated circuit packages).

The multi-layer ceramic chip carrier 972 is used to build high-Qinductors, transformers, and capacitors. This can be beneficial for CMOSpower amplifier architecture since multiple inductors and capacitors maybe required. For example, a single band power amplifier may require 4-8inductors which would be impractical to build on a printed circuitboard. In addition, multiple matching networks are used to provide thehigh transformation ratio required in a push-pull amplifier design. Inone embodiment of the invention, the transformers, inductors,capacitors, and other passive devices are formed on the ceramic chipcarrier 972. The ceramic chip carrier 972 includes multiple conductivelayers 978 (shown as hidden lines) that can be designed to implementthese passive devices.

In one embodiment of the RF power amplifier shown in FIG. 14, all of theswitching devices and capacitors C2+ and C2 reside on the integratedcircuit 970, with the inductors L3+, L3−, L5, L1+, L1−, L1−, L2+, andL2− residing on the multi-layer ceramic chip carrier 972.

In a CMOS power amplifier design, multiple high-Q inductors are requiredto tune out large on-chip gate capacitances. Since these capacitancesare large, the required inductors are low in value and difficult tointegrate. One solution is to form high-Q inductors on the ceramic chipcarrier. FIG. 27 is a diagram illustrating the ceramic chip carrier 972shown in FIGS. 25 and 26 with a horizontally-formed inductor 1180 formedin the ceramic chip carrier 972. The inductor 1180 includes a horizontalloop portion formed by conductive trace 1182 connected to two bumps 974of the ceramic chip carrier 972 by two vias 1184. One disadvantage withthe inductor 1180 is that the inductor connection points needs to beclose to the edge of the ceramic chip carrier 972 unless the value ofthe inductor is large enough to route to a lower layer of the ceramicchip carrier 972.

FIG. 28 is a diagram illustrating the ceramic chip carrier 972 with avertically-formed inductor 1280 formed in the carrier 972. The inductor1280 is formed in the vertical direction by vias 1284 extending toconductive trace 1286, which may be formed on a lower level of thecarrier 972. As shown, the inductor 1280 extends downward into theceramic chip carrier 972 and is coplanar, since the vias 1284 and trace1286 exist on the same plane. The vias 1284 may be formed throughseveral layers of the carrier 972, depending the inductance desired. Avertically-formed inductor such as the inductor 1280 has two majoradvantages over horizontally-formed inductors. First, thevertically-formed inductors can be formed underneath the chip 970without blocking other routing channels. Therefore, more layout optionsare available, and more inductors can be formed. Second, thevertically-formed vias 1284, as opposed to the horizontal conductivetrace 1182, result in less loss at RF frequencies since the viasl284have a greater cross-sectional surface area than the conductive traces.The vias 1284 are substantially cylindrical and have a surface area ofπdL, where d is the diameter of the via 1284 (e.g., 100 μm) and L is thelength of the via. The conductive traces, such as conductive trace 1182,have a surface area of 2 dL. Therefore, the resistance of a via at RFfrequencies is approximately π/2 less than the resistance of aconductive trace 1182.

FIGS. 29 and 30 illustrate one embodiment of vertically-formed inductorsof the present invention. FIG. 29 is an electrical schematic diagramshowing inductors L7, L8, L9, L10, and L11 connected between connectionpoints 1310, 1312, 1314, and 1316. As shown, inductors L7 and L8 areconnected between connection points 1310 and 1312. Similarly, inductorsL9 and L10 are connected between connection points 1314 and 1316.Inductor L11 is connected between connection points 1318 and 1320, whichare formed between inductors L9 and L10, and L7 and L8.

FIG. 30 illustrates an example of how the circuit of FIG. 29 can beimplemented using vertically-formed inductors of the present invention.The connection points 1310, 1312, 1314, and 1316 are formed at thesurface of the ceramic chip carrier (not shown in FIG. 30) and will beelectrically connected to four of the bumps 974 of the flip-chip 970. Inthis example, the inductors are formed using the upper two layers of theceramic chip carrier. Vias 1322 and 1324 extend through both layerswhere they are connected to an end of conductive traces 1326 and 1328,respectively, formed in the lower layer of the ceramic chip carrier. Theopposite ends of the conductive traces 1326 and 1328 are connected tovias 1330 and 1332, respectively, which are also formed in the lowerlayer of the ceramic chip carrier. Together, the via 1322, conductivetrace 1326, and via 1330 form inductor L7. Similarly, the via 1324,conductive trace 1328, and via 1332 form inductor L9. The vias 1330 and1332 are connected to opposite ends of conductive trace 1334, formed inthe upper layer. The conductive trace 1334 forms the inductor L11.Finally, vias 1336 and 1338 are connected to the vias 1330 and 1332,respectively, as well as to opposite ends of the conductive trace 1334.The vias 1336 and 1338 form the inductors L8 and L10, respectively.While FIGS. 29 and 30 show one specific example of how inductors couldbe formed in the ceramic chip carrier, it should be understood thatother implementations are possible.

In the preceding detailed description, the invention is described withreference to specific exemplary embodiments thereof. Variousmodifications and changes may be made thereto without departing from thebroader spirit and scope of the invention as set forth in the claims.The specification and drawings are, accordingly, to be regarded in anillustrative rather than a restrictive sense.

1. A power amplifier comprising: a first switching device connectedbetween a first supply voltage and a first output node; a secondswitching device connected between a second supply voltage and a secondoutput node; and an inductance coupled between the first and secondoutput nodes.
 2. The power amplifier of claim 1, wherein a load iscoupled to the second output node.
 3. The power amplifier of claim 1,further comprising a first capacitor coupled to the first output node,and a second capacitor coupled to the second output node.
 4. The poweramplifier of claim 1, further comprising a first capacitor coupled tothe first output node.
 5. The power amplifier of claim 1, furthercomprising a first capacitor coupled between the first and second outputnodes.
 6. The power amplifier of claim 1, wherein the first and secondswitching devices are driven by signals that repeatedly turn the deviceson and off.
 7. The power amplifier of claim 6, wherein the first andsecond switching devices are both cycled on during the same time period,and wherein the first and second switching devices are both cycled offduring the same time period.
 8. The power amplifier of claim 4, whereinthe first capacitor is provided by the input capacitance of a thirdswitching device.
 9. The power amplifier of claim 3, wherein the firstcapacitor is provided by the input capacitance of a third switchingdevice, and wherein the second capacitor is provided by the inputcapacitance of a fourth switching device.
 10. The power amplifier ofclaim 1, wherein the first and second switching devices are comprised oftransistors.
 11. The power amplifier of claim 1, wherein the firstswitching device is comprised of a PMOS transistor, and wherein thesecond switching device is comprised of an NMOS transistor.
 12. Thepower amplifier of claim 1, further comprising a load coupled across thefirst and second output nodes.
 13. The power amplifier of claim 1,further comprising a transformation network coupled to the first andsecond output nodes.
 14. The power amplifier of claim 13, wherein thetransformation network further comprises: a capacitor coupled to thefirst output node and a third node; an inductor coupled to the secondoutput node and the third node; and a load coupled to the third node.15. The power amplifier of claim 1, further comprising a preamplifierconnected to the power amplifier, the preamplifier further comprising: athird switching device connected between said first supply voltage and athird node and coupled to the input to the first switching device; afourth switching device connected between said second supply voltage anda fourth node and coupled to the input to the second switching device;and a second inductor coupled between the third and fourth nodes. 16.The power amplifier of claim 1, further comprising one or more inductorscoupled between the first output node and a third supply voltage. 17.The power amplifier of claim 1, further comprising one or more inductorscoupled between the first output node and a third supply voltage, andone or more inductors coupled between the second output node and afourth supply voltage.
 18. A method of reducing the peak output voltageof an amplifier comprising the steps of: providing an inductor havingfirst and second terminals; providing a first switching device connectedbetween the first terminal of the inductor and a first supply voltage;providing a second switching device connected between the secondterminal of the inductor and a second supply voltage; applying a voltagebetween the first and second terminals of the inductor during a firstportion of a clock cycle by turning on the first and second switchingdevices; and turning off the first and second switching devices during asecond portion of the clock cycle.
 19. The method of claim 18, furthercomprising the steps of providing a first capacitance connected to thefirst terminal, providing a second capacitance connected to the secondterminal, wherein current from the inductor charges or discharges thefirst and second capacitances during the second portion of the clockcycle.
 20. The method of claim 18, further comprising the step ofconnecting a load to the first node. 21-46. (canceled).